Communication Concepts AN758 - Manual (Page 5)

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HIVE INF
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RMA
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PRODUCT TRANSFERRED T
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M/A
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COM
AN758
5 RF Application Reports
Although omitted from the preliminary calculations, the
2 x 5 nH inductances, comprising of lead length, were in-
cluded in this program.
The input transformer is a 9:1 type, and uses a television
antenna balun type ferrite core, made of high permeability
material. The low impedance winding consists of one turn
of 1/8
″ copper braid. The sections going through the
openings in the ferrite core are rounded to resemble two
pieces of tubing electrically. The primary consists of AWG
#22 TFE insulated wire, threaded through the rounded
sections of braid, placing the primary and secondary leads
in opposite ends of the core.
(4) (5) The saturation flux density
is about 60 gauss which is well below the limits for this core.
For calculation procedures, see discussion about the output
transformer.
This type physical arrangement provides a tight coupling,
reducing the amount of leakage flux at high frequencies. The
wire gauge, insulation thickness, and number of strands have
a minimal effect in the performance except at very high
impedance ratios, such as 25:1 and up. The transformer
configuration is shown in Figure 4. By using a vector
impedance meter, the values for C3 and C4 were measured
to give a reasonable input match at 30 MHz, (Z
in
= 1.62
– j 0.21 x 2 = 3.24 – j 0.42) with the smallest possible phase
angle.
SECONDARY C4 C3 50 Ω 470 pF 56 pF
Figure 4. Transformer Configuration
When the high impedance side was terminated into 50
Ω, the following readings were obtained at the secondary:
The VSWR was calculated as
Z1 + Z2
Z1 – Z2
where:
Z1 = Impedance at transformer secondary.
Z2 = Input impedance of compensation network x 2 (R
S
in
Figures 2 and 3) as in computer data presented
ahead.
The effect of the lower VSWR to the power loss in the input
network can be calculated as follows:
10 Log
1 –
S1 + 1
S1 – 1
Ǔ ǒ
2
1 –
S2 + 1
S2 – 1
Ǔ ǒ
2
where:
S1 = VSWR 1 (Lower)
S2 = VSWR 2 (Higher)
which at 30 MHz = 10 Log
1 –
1.11 + 1
1.11 – 1
Ǔ ǒ
2
1 –
1.74 + 1
1.74 – 1
Ǔ ǒ
2
= 0.32 dB, 2.7 – 0.32 = 2.38 dB
0.927
0.997
Ǔ ǒ
= 10 Log
These figures for other frequencies are presented with the
data below. Later, some practical experiments were done
with moving the resonance of C5 L5 lower, to find out if insta-
bilities would occur in a practical circuit. When the resonance
was equal to the test frequency, slight breakup was noticed
in the peaks of a two-tone pattern. It was then decided to ad-
just the resonance to 31 MHz, where C5 = 560 pF, and the
phase angle at 30 MHz increases to 87
°. The transducer loss
is further reduced by about 0.2 dB.
Table 2.
Frequency
MHz
R
S
Ohms
X
S
Ohms
VSWR Attenuation
dB
2.0 5.59 + 0.095 1.05 12.99
4.0 5.55 + 0.057 1.15 12.06
7.5 5.50 + 0.046 1.32 10.40
15 4.90 + 0.25 1.48 7.28
20 4.32 + 0.55 1.38 5.63
30 3.43 + 0.73 1.11 2.38
* Above readings with transformer and compensation network.
Several types of output transformer configurations were
considered. The 12.5 Q collector-to-collector impedance
estimated earlier, would require a 4:1 transformer for a 50
Ω
output. The type used here as the input transformer exhibits
good broad band characteristics with a convenient physical
design. However, according to the low frequency minimum
inductance formula presented earlier in connection with T2,
the initial permeability required would be nearly 3000, with
the largest standard core size available. High permeability
ferrites are almost exclusively of Nickel-Manganese
composition, and are lossy at radio frequencies. Although
their Curie points are higher than those of lower permeability
Nickel-Zinc ferrites, the core losses would degrade the
amplifier performance. With the core losses being a function
of the power level, these rules can sometimes be
disregarded in low power applications.
A coaxial cable version was adapted for this design, since
the transmission line type transformers are theoretically ideal
for RF applications, especially in the 1:4 impedance ratio.
A balanced to unbalanced function would normally require
three separate transmission lines including a balun
(5) (6) . It
appears that the third line can be omitted, if lines a and b
(Figure 3) are wound on separate magnetic cores, and the
physical length of the lines is sufficient to provide the
necessary isolation between the collectors and the load. In
accordance to formulas in
(7) , the minimum line length
required at 2 MHz, employing Stackpole 57-9074 or
equivalent ferrite toroids is 4.2
″, and the maximum
permissible line length at 30 MHz would be approximately
20
″. The 4.2″ amounts to four turns on the toroid, and
measures 1.0 µH, which in series with the second line is
sufficient for 2 MHz. Increasing the minimum required line
length by a factor of 4 is still within the calculated limits, and
in practical measurements the isolation has been found to
be over 30 dB across the band. The main advantage with
this arrangement is a simplified electrical and physical
lay-out.
The maximum flux density of the toroids is approximately
200 gauss (3) , and the number of turns has been increased
beyond the point where the flux density of the magnetic core
is the power limiting factor.